Electronic converters for lighting sources comprising, for example, at least one LED (Light Emitting Diode) or other solid-state lighting means may provide a direct current output. Such current may be stable or vary in time, for example in order to regulate the light intensity emitted by the lighting source (so called dimming function).
FIG. 1 shows a possible lighting system comprising an electronic converter 10 and a lighting module 20, comprising for example at least one LED L.
Electronic converter 10 usually comprises a control circuit 102 and a power circuit 12 (for example an AC/DC or DC/DC switching supply) and provides as output a direct current through a power output 106. Such a current may be stable, or else may vary in time. For example, control circuit 102 may set, via a reference channel Iref of power circuit 12, the current required by LED module 20.
For example, such a reference channel Iref may be used in order to regulate the intensity of the light emitted by lighting module 20. Actually, a regulation of the light intensity emitted by LED module 20 may be generally achieved by regulating the average current flowing through lighting module 20, for example by setting a lower reference current Iref, or by switching on or off power circuit 12 through a Pulse Width Modulation (PWM) signal.
However, the case wherein module 20 is supplied with a regulated voltage, i.e. wherein converter 12 is a voltage generator, typically requires a current regulator which is connected in series with lighting sources L, in order to limit the current. In this case, the dimming function may be implemented also via such a current regulator, for example:
a) by selectively switching on or off such a current regulator via a driving signal, e.g. a PWM signal, or
b) in case of an adjustable current regulator, by setting the reference current of such a current regulator.
Generally speaking there are many types of electronic converters, which are divided mainly into isolated and non-isolated converters. For example, non-isolated electronic converters are “buck”, “boost”, “buck-boost”, “Cuk”, “SEPIC” and “ZETA” converters. On the contrary, isolated converters are for example “flyback”, “forward”, “Half-bridge” and “Full-bridge” converters. Such kinds of converters are well known to the skilled in the art.
For example, FIG. 2 shows the circuit arrangement of a flyback converter.
As is well-known, a flyback converter comprises a transformer T with a primary winding T1 and a secondary winding T2, an electronic switch S, such as for example an n-channel MOSFET transistor (Metal-Oxide-Semiconductor Field-Effect Transistor), or a bipolar or IGBT transistor (Insulated-Gate Bipolar Transistor), a diode D1 and an output capacitor Co.
Specifically, transformer T may be modelled as an inductance Lm, connected in parallel with primary winding T1, which represents the magnetising inductance of transformer T, and an ideal transformer with a given turn ratio 1:n.
In the presently considered example, converter 12 receives as input, via two input terminals, a voltage Vin, and provides as output, via a supply line 106, a regulated current iout. Those skilled in the art will appreciate that voltage Vin may also be obtained through an input AC current, for example via a diode or a diode bridge rectifier, and optionally a filtering capacitor.
Specifically, the first input terminal is connected to the first terminal of primary winding T1 of transformer T and the second input terminal represents a first ground GND1. On the contrary, the second terminal of primary winding T1 of transformer T is connected through switch S to ground GND1. Therefore, switch S may be used to selectively activate the current flow through primary winding T1 of transformer T.
On the other hand, the first terminal of secondary winding T2 of transformer T is connected through a diode D1 to a first output terminal, which represents power output 106, and the second terminal of secondary winding T2 of transformer T is connected directly to a second output terminal, which represents a second ground GND2, which due to the isolating effect of transformer T is preferably different from ground GND1 and is therefore denoted with a different ground symbol.
Finally, an output capacitor Co is connected in parallel with the output, i.e. between terminals 106 and GND2.
Therefore, when switch S is closed, primary winding T1 of transformer T is connected directly to input voltage Vin. This causes an increase of the magnetic flow in transformer T. Therefore, the voltage across secondary winding T2 is negative, and diode D1 is inversely biased. In this condition, output capacitor Co provides the energy required by the load, for example by lighting module 20.
On the contrary, when switch S is open, the energy stored in transformer T is transferred as flyback current to lighting module 20.
As previously mentioned, the control may be in current or voltage. To this purpose, a control unit 112 is typically used which drives switch S so that output voltage Vout or output current iout is regulated on a desired value, so as for example reference current Iref. To this purpose it is possible to use, as known in itself, a sensor adapted to detect current iout or voltage Vout.
Typically, control unit 112 drives switch S with Pulse Width Modulation (PWM), wherein switch S is closed during a first operation interval and switch S is opened during a second operation interval. Those skilled in the art will appreciate that such PWM driving and the control of duration of operation intervals are well known, and may be implemented, for example, through a feedback of the output voltage or current through an error amplifier. For example, in the case of a current control, the duration of the first interval is increased until the (average) output current reaches a predetermined threshold.
Such a PWM driving may involve three different operation modes. Specifically, if the current in the magnetising inductance Lm never reaches zero during a switching cycle, the converter is said to operate in a Continuous Current Mode (CCM). On the contrary, when the current in the magnetising inductance Lm reaches zero during the period, the converter is said to operate in a Discontinuous Current Mode (DCM). Typically, the converter operates in a discontinuous mode when the load absorbs a low current, and in a continuous mode at higher levels of absorbed current. The border between the continuous mode, CCM, and the discontinuous mode, DCM, is reached when the current reaches zero, exactly at the end of the switching cycle. Such a limit case is referred to as Transition Mode (TM). Moreover, there is the possibility of driving the switch with a resonant or quasi-resonant driving, wherein switch S is switched when the voltage across said electronic switch (S) is zero, or when a local minimum is reached. Typically the switching frequency, i.e. the sum of the duration of operation periods, is fixed for a CCM or a DCM driving, and is variable for a quasi-resonant driving.
However a flyback converter, and generally every switching power supply, comprises parasitic elements. For example, in a flyback converter one of the most influential elements is transformer T, particularly its leakage inductance. For example, in FIG. 2, the leakage inductance of transformer T is modelled as an inductance Lr connected in series with the secondary winding T2 of transformer T. In a similar way, in a forward converter, both the magnetizing inductance Lm and the leakage inductance Lr constitute parasitic elements. Actually such inductances store energy which often cannot be transferred to the load. For example, in a flyback converter the discharge of the energy stored in the parasitic inductance Lr may cause an overvoltage across switch S. Moreover the zeroing of current through switch S cannot take place with zero voltage, which involves switching losses as well.
Therefore snubbers have been used in the past. Such snubbers are typically divided into the following categories:                dissipative snubber: a dissipative network comprising passive components, particularly resistors;        non-dissipative passive snubber: a circuit comprising one or several reactive components, for example capacitors, which allow for the recovery of the energy stored in the inductive components; and        non dissipative active snubber: a circuit comprising a passive network and one or several switches.        
Snubber circuits also have other advantages, such as:                Electromagnetic Interference (EMI) is typically reduced; and        the switching of the switch or switches of the switching supply may take place at zero voltage: it is the so-called Zero-Voltage Switching (ZVS).        
Details on the operation of passive snubber circuits are described, for example, in P. C. Todd, “Snubber circuits: Theory, Design and Application”, Unitrode Corporation, May 1993, the content whereof is incorporated herein by reference.
Details on the operation of non-dissipative snubber circuits, for example for flyback converters, are described in T. Ninomiya, T. Tanaka, and K. Harada, “Analysis and optimization of a non-dissipative LC turn-off snubber,” IEEE Transactions on Power Electronics, vol. 3, no. 2, pp. 147-156, 1988, or in Chih-Sheng Liao, Keyue M. Smedley, “Design of high efficiency Flyback converter with energy regenerative snubber”, Conference: Applied Power Electronics Conference and Exposition Annual IEEE Conference—APEC, pp. 796-800, 2008, the contents whereof are incorporated herein by reference.
Finally, active snubber circuits are described for example in B. Andreycak, “Active clamp and reset technique enhances forward converter performance”, Unitrode Power Supply Design Seminar, SEM-1000, pp. 3-1-3-18, 1994 for forward converters, and in Robert Watson, et al., “Utilization of an Active-Clamp Circuit to Achieve Soft Switching in Flyback converters”, IEEE Transactions on Power Electronics, V. 11, pp. 162-169, 1996 for flyback converters, the contents whereof are incorporated herein by reference.
The previously described snubber circuits have common features, as they are located on the primary side of the transformer, and they limit the peak or rise-time of the voltage across the main switch. Therefore, such circuits cannot directly snub effects which are caused by components on the secondary side of the transformer.
Document EP 1 202 440 A1 discloses a snubber circuit which is located at the secondary side of a transformer. Specifically, the disclosed snubber circuit comprises two diodes, a capacitor and an inductor. According to document EP 1 202 440 A1 the inductor of such a snubber circuit permits that the current of the primary side switch is smoothly and gradually decreased due to the current supplied from the capacitor of the snubber circuit, so that the voltage of the primary side switch increases with a gradient that a voltage ringing is suppressed.
The inventor has noted that the arrangements disclosed in document EP 1 202 440 A1 have several inconveniences. For example, according to this document, the snubber capacitor on the secondary side is discharged with the same current that will supply the load. Thus, in order to have an important slope reduction of the rise time of the voltage of the primary side switch, a high capacitance is required for the snubber capacitor. However, a high capacitance value implies that also a high inductance is required for the inductor of the snubber circuit in order to charge the snubber capacitor during the switch-on time. Moreover, when the output diode starts to conduct, the leakage inductance of the transformer and the primary side switch capacitance will tend to oscillate as is shown e.g. in FIG. 20A of document EP 1 202 440 A1.